- •Analysis and Application of Analog Electronic Circuits to Biomedical Instrumentation
- •Dedication
- •Preface
- •Reader Background
- •Rationale
- •Description of the Chapters
- •Features
- •The Author
- •Table of Contents
- •1.1 Introduction
- •1.2 Sources of Endogenous Bioelectric Signals
- •1.3 Nerve Action Potentials
- •1.4 Muscle Action Potentials
- •1.4.1 Introduction
- •1.4.2 The Origin of EMGs
- •1.5 The Electrocardiogram
- •1.5.1 Introduction
- •1.6 Other Biopotentials
- •1.6.1 Introduction
- •1.6.2 EEGs
- •1.6.3 Other Body Surface Potentials
- •1.7 Discussion
- •1.8 Electrical Properties of Bioelectrodes
- •1.9 Exogenous Bioelectric Signals
- •1.10 Chapter Summary
- •2.1 Introduction
- •2.2.1 Introduction
- •2.2.4 Schottky Diodes
- •2.3.1 Introduction
- •2.4.1 Introduction
- •2.5.1 Introduction
- •2.5.5 Broadbanding Strategies
- •2.6 Photons, Photodiodes, Photoconductors, LEDs, and Laser Diodes
- •2.6.1 Introduction
- •2.6.2 PIN Photodiodes
- •2.6.3 Avalanche Photodiodes
- •2.6.4 Signal Conditioning Circuits for Photodiodes
- •2.6.5 Photoconductors
- •2.6.6 LEDs
- •2.6.7 Laser Diodes
- •2.7 Chapter Summary
- •Home Problems
- •3.1 Introduction
- •3.2 DA Circuit Architecture
- •3.4 CM and DM Gain of Simple DA Stages at High Frequencies
- •3.4.1 Introduction
- •3.5 Input Resistance of Simple Transistor DAs
- •3.7 How Op Amps Can Be Used To Make DAs for Medical Applications
- •3.7.1 Introduction
- •3.8 Chapter Summary
- •Home Problems
- •4.1 Introduction
- •4.3 Some Effects of Negative Voltage Feedback
- •4.3.1 Reduction of Output Resistance
- •4.3.2 Reduction of Total Harmonic Distortion
- •4.3.4 Decrease in Gain Sensitivity
- •4.4 Effects of Negative Current Feedback
- •4.5 Positive Voltage Feedback
- •4.5.1 Introduction
- •4.6 Chapter Summary
- •Home Problems
- •5.1 Introduction
- •5.2.1 Introduction
- •5.2.2 Bode Plots
- •5.5.1 Introduction
- •5.5.3 The Wien Bridge Oscillator
- •5.6 Chapter Summary
- •Home Problems
- •6.1 Ideal Op Amps
- •6.1.1 Introduction
- •6.1.2 Properties of Ideal OP Amps
- •6.1.3 Some Examples of OP Amp Circuits Analyzed Using IOAs
- •6.2 Practical Op Amps
- •6.2.1 Introduction
- •6.2.2 Functional Categories of Real Op Amps
- •6.3.1 The GBWP of an Inverting Summer
- •6.4.3 Limitations of CFOAs
- •6.5 Voltage Comparators
- •6.5.1 Introduction
- •6.5.2. Applications of Voltage Comparators
- •6.5.3 Discussion
- •6.6 Some Applications of Op Amps in Biomedicine
- •6.6.1 Introduction
- •6.6.2 Analog Integrators and Differentiators
- •6.7 Chapter Summary
- •Home Problems
- •7.1 Introduction
- •7.2 Types of Analog Active Filters
- •7.2.1 Introduction
- •7.2.3 Biquad Active Filters
- •7.2.4 Generalized Impedance Converter AFs
- •7.3 Electronically Tunable AFs
- •7.3.1 Introduction
- •7.3.3 Use of Digitally Controlled Potentiometers To Tune a Sallen and Key LPF
- •7.5 Chapter Summary
- •7.5.1 Active Filters
- •7.5.2 Choice of AF Components
- •Home Problems
- •8.1 Introduction
- •8.2 Instrumentation Amps
- •8.3 Medical Isolation Amps
- •8.3.1 Introduction
- •8.3.3 A Prototype Magnetic IsoA
- •8.4.1 Introduction
- •8.6 Chapter Summary
- •9.1 Introduction
- •9.2 Descriptors of Random Noise in Biomedical Measurement Systems
- •9.2.1 Introduction
- •9.2.2 The Probability Density Function
- •9.2.3 The Power Density Spectrum
- •9.2.4 Sources of Random Noise in Signal Conditioning Systems
- •9.2.4.1 Noise from Resistors
- •9.2.4.3 Noise in JFETs
- •9.2.4.4 Noise in BJTs
- •9.3 Propagation of Noise through LTI Filters
- •9.4.2 Spot Noise Factor and Figure
- •9.5.1 Introduction
- •9.6.1 Introduction
- •9.7 Effect of Feedback on Noise
- •9.7.1 Introduction
- •9.8.1 Introduction
- •9.8.2 Calculation of the Minimum Resolvable AC Input Voltage to a Noisy Op Amp
- •9.8.5.1 Introduction
- •9.8.5.2 Bridge Sensitivity Calculations
- •9.8.7.1 Introduction
- •9.8.7.2 Analysis of SNR Improvement by Averaging
- •9.8.7.3 Discussion
- •9.10.1 Introduction
- •9.11 Chapter Summary
- •Home Problems
- •10.1 Introduction
- •10.2 Aliasing and the Sampling Theorem
- •10.2.1 Introduction
- •10.2.2 The Sampling Theorem
- •10.3 Digital-to-Analog Converters (DACs)
- •10.3.1 Introduction
- •10.3.2 DAC Designs
- •10.3.3 Static and Dynamic Characteristics of DACs
- •10.4 Hold Circuits
- •10.5 Analog-to-Digital Converters (ADCs)
- •10.5.1 Introduction
- •10.5.2 The Tracking (Servo) ADC
- •10.5.3 The Successive Approximation ADC
- •10.5.4 Integrating Converters
- •10.5.5 Flash Converters
- •10.6 Quantization Noise
- •10.7 Chapter Summary
- •Home Problems
- •11.1 Introduction
- •11.2 Modulation of a Sinusoidal Carrier Viewed in the Frequency Domain
- •11.3 Implementation of AM
- •11.3.1 Introduction
- •11.3.2 Some Amplitude Modulation Circuits
- •11.4 Generation of Phase and Frequency Modulation
- •11.4.1 Introduction
- •11.4.3 Integral Pulse Frequency Modulation as a Means of Frequency Modulation
- •11.5 Demodulation of Modulated Sinusoidal Carriers
- •11.5.1 Introduction
- •11.5.2 Detection of AM
- •11.5.3 Detection of FM Signals
- •11.5.4 Demodulation of DSBSCM Signals
- •11.6 Modulation and Demodulation of Digital Carriers
- •11.6.1 Introduction
- •11.6.2 Delta Modulation
- •11.7 Chapter Summary
- •Home Problems
- •12.1 Introduction
- •12.2.1 Introduction
- •12.2.2 The Analog Multiplier/LPF PSR
- •12.2.3 The Switched Op Amp PSR
- •12.2.4 The Chopper PSR
- •12.2.5 The Balanced Diode Bridge PSR
- •12.3 Phase Detectors
- •12.3.1 Introduction
- •12.3.2 The Analog Multiplier Phase Detector
- •12.3.3 Digital Phase Detectors
- •12.4 Voltage and Current-Controlled Oscillators
- •12.4.1 Introduction
- •12.4.2 An Analog VCO
- •12.4.3 Switched Integrating Capacitor VCOs
- •12.4.6 Summary
- •12.5 Phase-Locked Loops
- •12.5.1 Introduction
- •12.5.2 PLL Components
- •12.5.3 PLL Applications in Biomedicine
- •12.5.4 Discussion
- •12.6 True RMS Converters
- •12.6.1 Introduction
- •12.6.2 True RMS Circuits
- •12.7 IC Thermometers
- •12.7.1 Introduction
- •12.7.2 IC Temperature Transducers
- •12.8 Instrumentation Systems
- •12.8.1 Introduction
- •12.8.5 Respiratory Acoustic Impedance Measurement System
- •12.9 Chapter Summary
- •References
General Properties of Electronic Single-Loop Feedback Systems |
175 |
clear that four major types of feedback exist: (1) negative voltage feedback;
(2) negative current feedback; (3) positive voltage feedback; and (4) positive current feedback. Negative voltage feedback is most widely encountered in electronic engineering practice. Its properties will be examined in the following sections.
4.3Some Effects of Negative Voltage Feedback
4.3.1Reduction of Output Resistance
Figure 4.2 illustrates a simple Thevenin amplifier circuit with NVB applied to its summing junction (input node). Assume that the amplifier’s input resistor, Rin, RF and Rs, so Rin can be ignored in writing the node equation for Vi. Also, in finding the NFB amplifier’s open-circuit voltage gain, assume that Ro RF, so negligible voltage drop takes place across Ro. Thus, the node equation is:
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Vi = −Vo/Kv |
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FIGURE 4.2
Schematic of a simple Thevenin VCVS with negative feedback. See text for analysis.
© 2004 by CRC Press LLC
176 Analysis and Application of Analog Electronic Circuits
Thus, Equation 4.6 can be used to write the closed-loop system’s transfer
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AL = −Kv [Rs/(RF + Rs)] (Note: system has NFB.) |
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In op amps, Kv is numerically very large, so Equation 4.7 reduces to the well-known gain relation for the ideal op amp inverter: Vo/Vs = −RF/Rs.)
There are two common ways to find the output resistance of the simple NFB amplifier of Figure 4.2. One is to set Vs = 0, then place an independent test source, Vt, between the Vo node and ground and calculate Rout = Vt/It. The other method is to measure the open-circuit output voltage using Equation 4.7, then calculate the short-circuit output current (with Vo = 0). Clearly, Rout = VOC/IoSC. Inspection of Figure 4.2 with a shorted output lets one write:
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In general, the NFB amplifier’s output impedance can be written as a function of frequency:
Zout(jω) = Ro/RD(jω) |
(4.13) |
Note that, in general, Zout(jω) is very small (e.g., milliohms) at dc and low frequencies, then gradually increases with ω. This property can be demonstrated by assuming Kv to have a frequency response of the form:
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© 2004 by CRC Press LLC
General Properties of Electronic Single-Loop Feedback Systems |
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Equation now yields: |
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From the preceding equation it is clear that Zo Ro/βKvo at dc, increasing with frequency to Zo = Ro at frequencies above βKvo/τ r/s.
4.3.2Reduction of Total Harmonic Distortion
A very important property of NFB is the reduction of total harmonic distortion (THD) at the output of amplifiers, in particular power amplifiers. First it is necessary to examine what THD means. Assume that a certain amplifier has a soft saturation nonlinearity modeled by:
y = ax + bx2 + cx3 + dx4 + ex5 +… |
(4.16) |
Clearly, a = Kv, the linear gain. The b, c, d, e, … terms can be zero or have either sign. They give rise to the harmonic distortion that can be the result of transistor cut-off or saturation, or power supply saturation. If x(t) = Xo sin(ωot), the output y(t) will contain the desired fundamental frequency from the a-term, as well as unwanted harmonics. For example,
y(t) = a Xo sin(ωot) + (bXo2 2)[1 − cos(2ωot)]+ (cXo3 4)
(4.17)
[3 sin(ωot) − sin(3ωot)]+ ≡
The power terms approximating the nonlinear transfer characteristic of the amplifier generate dc, fundamental, and harmonics of the order of the nonlinear term in addition to the desired fundamental frequency. These can be summarized by the expression:
y(t) = Kv Xo sin(ωot) + H2 sin(2ωot) + H3 sin(3ωot) + H4 sin(4ωot) +… (4.18)
The total RMS harmonic distortion is defined as:
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THD = H22 2 + H32 2 + H42 2 + ≡ = Hk2 2 |
(4.19) |
k=2
Note that Hk2/2 is the mean squared value of the kth harmonic; (Kv2Xo2/2) is the MS value of the fundamental frequency (desired) component of the output. In general, the larger Xo is, the larger will be the THD.
© 2004 by CRC Press LLC
178 |
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Analysis and Application of Analog Electronic Circuits |
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NFB System with Output Distortion |
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THD (rmsV)
Harmonic Generation
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FIGURE 4.3
(A)Block diagram of a SISO negative voltage feedback system with output harmonic distortion voltage introduced in the last (power) stage. A sinusoidal input of frequency ωo is assumed.
(B)Rms power spectrum of the feedback amplifier output. vos1 is the RMS amplitude of the fundamental frequency output. (C) Plot of how total harmonic distortion typically varies as a function of the amplitude of the fundamental output voltage.
Figure 4.3(A) illustrates a block diagram showing how the RMS THD is added to the output of an NFB amplifier. Distortion is assumed to occur in the last stage. If there were no feedback, (β = 0), Vo = Vs αKv + vd (VsαKv). Vs is adjusted so that the RMS output signal is vos1. With feedback,
© 2004 by CRC Press LLC
General Properties of Electronic Single-Loop Feedback Systems |
179 |
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With NFB, the output THD is reduced by a factor of 1/(1 + β Kv), given that the signal output is the same in the nonfeedback and the feedback cases. This result has vast implications in the design of nearly distortion-free audio power amplifiers, as well as linear signal conditioning systems in general that use op amps. vdk in Figure 4.3(B) is simply the kth RMS harmonic voltage; i.e., vdk = Hk/ 2. Figure 4.3(C) shows how the RMS THD increases as the signal output is made larger. At vos = VCL, the level of distortion increases abruptly because the amplifier begins to saturate and clip the output signal waveform.
In good low-distortion amplifier design, the input stages are made as linear as possible, and the amplifier is designed so that the output (power) stage saturates before the input and intermediate gain stages do. This allows the THD voltage to effectively be added in the last stage, and thus the THD is reduced by a factor of 1/(1 − AL) = 1/RD.
4.3.3Increase of NFB Amplifier Bandwidth at the Cost of Gain
It is axiomatic in the design of electronic circuit that negative feedback allows one to extend closed-loop amplifier bandwidth (BW) at the expense of dc or mid-band gain. Another way of looking at the effect of NFB on BW is to observe that every amplifier is endowed with some gain*bandwidth product (GBWP); as a figure of merit, larger is better. GBWP remains relatively constant as feedback is applied, so when NFB reduces gain, it increases BW. The frequency-compensated (FC) op amp makes an excellent example of GBWP constancy. The open-loop gain of an FC op amp can be given by:
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Note that the op amp is a difference amplifier and has a gain*bandwidth product:
GBWPoa = Kvo/(2πτa) Hertz |
(4.22) |
Now connect the OA as a simple noninverting amplifier, as shown in Figure 4.4(A). The amplifier’s output is:
Vo = (Vs − βVo)Kvo/(sτa + 1) |
(4.23) |
© 2004 by CRC Press LLC
180 |
Analysis and Application of Analog Electronic Circuits |
RF
R1 Vi’
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B
FIGURE 4.4
(A) An op amp connected as a noninverting amplifier. (B) An op amp connected as an inverting amplifier.
where
β ∫ R1/(R1 + RF)
Solving for the voltage gain yields:
Vo |
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(1+ βKvo ) |
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The GBWP for this closed-loop amplifier is simply:
GBWPamp = |
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(4.24)
(4.25)
(4.26)
For the closed-loop feedback amplifier to have exactly the same GBWP as the open-loop op amp is a special case. Note that the closed-loop gain is divided by the RD and the closed-loop BW is multiplied by the RD.
Next, consider the gain, BW, and GBWP of a simple op amp inverter, shown in Figure 4.4(B). To attack this problem, write a node equation on the summing junction (Vi′) node:
Vi′(G1 + GF) − Vo GF = Vs G1 |
(4.27) |
© 2004 by CRC Press LLC