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III.

There are two operating conditions giving rise to destruction via secondary breakdown. One is identified as FBSOA (forward-biased safe operating area). The other is identified as RBSOA (reverse-biased safe operating area). The SOA curves are graphical plots of collector current vs. collector voltage; they show what values of collector current and voltage are permissible, that is safe when applied simultaneously under prescribed conditions of temperature, bias, and pulse duration. Furthermore, the SOA curves tell you that heed must be paid not only to limits imposed by current, voltage, and power dissipation, but by secondary-breakdown energy constraints. Inasmuch as secondary breakdown is energy dependent, it should come as no surprise that secondary breakdown can occur at a lower collector voltage than that attributed to primary breakdown. In any event, the load line must not penetrate the SOA curves pertinent to the pulse duration, reverse bias, and to the junction temperature. Snubbed circuits can modify the load line in a favorable manner.

From a practical standpoint, a simple RC (resistive-capacitive) snubbed connected across the collector-emitter terminals of the switching transistor can be quite effective in absorbing the energy of switching transients, which might otherwise damage the transistor via reverse-bias secondary breakdown. For many applications, a 300 ohms, 0.02 uF (microfarad) combination is a good starting point. The resistance should be no inductive and be capable of dissipating 10 W. From this combination, it is usually possible to optimize final RC values. The procedure is to monitor the collector-emitter switching wave, paying heed to both fall time and the switching transient tends to be oscillator, even if the load is essentially resistive. The objective is to find an RC combination in the snubber that exerts the greatest damping of the switching transient, while minimally affecting the turn-off time of the voltage wave.

The energy dissipated in the resistive element of the RC snubbed must be supplied by the switching transistor during its forward-biased interval and will lower the operating efficiency of the switching circuit. These, however, are considered worthwhile trade offs; the transistor is more electrically rugged while forward biased, and the degradation in efficiency need not be excessive if the circuit is not over snubbed. In any event, a skimpy heat sink would be counterproductive in efforts to operate the transistor within its SOA boundaries. Also, care should be taken that transients on the ac utility line do not appear on the dc supply; these, sometimes, can be the straw that breaks the camel's back. (2687)

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TEXT 8

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Transient protection for bipolar switching transistors

Often, the terms snubber, damper, and clamp are used interchangeably. Although dampers and clamps limit load-line excursions, slumbers exert more modification of the entire load line. Protective circuit called a damper is much used in the horizontal output stage of TV sets.

All of these protective circuits share the common feature that, when properly deployed, destruction by secondary breakdown can be avoided. Such destruction otherwise tends to occur in switching applications when the energy stored in load or circuit inductance penetrates the RBSOA boundary of the transistor when it is turned off. Protection is accomplished via absorption of much of this excess energy.

Snubbing networks are nearly always used with inductive loads such as motors and solenoids. The inclusion of series resistance R, can provide additional dissipation of excess energy when the transistor is turned off. Over a wide range, the amount of inductance in the switching circuit is not the governing factor of RBSOA vulnerability to destruction; rather, it is the time constant of the inductive circuit, given by L/Rs ,. Note that the insertion of R, reduces this time constant. R, can also be viewed as a Q spoiler of the resonant circuit set up between L and circuit stray capacitance. High Q resonant circuits are reservoirs of high values of circulating energy; it is of course, preferable to dissipate LC (inductive-capacitive) energy in a resistance than in the collector junction of the transistor. In many prac­tical circuits, there is often sufficient resistance in the inductive load itself to serve this function.

In circuit C, an appropriately selected zener diode protects the transistor by clamping the collector-emitter voltage at, or below Vcex(sus). Although all zener diodes are fast acting, it is preferable to use zener diodes specially made for absorption of transient energy, such as Motorola's line of Mosorbs. Circuit G is commonly encountered in automotive ignition systems. Here, FBSOA is worsened because RBSOA energy regeneratively extends the on time of the transistor. That is considered a worthwhile trade off because transistors tend to be more electrically rugged in their FBSOA modes than when forced to absorb RBSOA energy. Some of the excess RBSOA energy is dissipated as heat in resistance R. (1951)

ТЕКСТ 9

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Switching transients in power MOSFET circuits

When power MOSFETs are used in switching applications, a particularly advanta­geous feature is that they, unlike bipolar transistors, are not susceptible to second­ary breakdown destruction.. It can be readily seen that the load-line limits imposed by the shaded area do not exist for the power MOSFET. This is especially rewarding in the high-voltage, low-current operational region where the bipolar transistor exhibits high vulnerability to secondary breakdown. Inasmuch as neither FBSOA or RBSOA boundaries are stipulated for the power MOSFET, it is often considered the more rugged and more forgiving switching device. However, certain precautions still must be observed, and it is common to incorporate similar protective circuits to those described for use in bipolar transistor switching applications.

At the voltage boundary an avalanche condition takes place. However, it is the better part of wisdom to avoid drain voltage avalanche in the power MOSFET. That is why snubbed and clamping tech­niques are often used. Even though it is easier to operate power MOSFETs safely without such protective networks than bipolar transistors, protection should at least be used during the experimental and bread boarding phases of circuit development.

A significant point of difference between MOS and bipolar devices is the vulnerability of the former to destruction of the gate by static charge during handling, or by voltage transients during operation. During experiments at least, it is a good idea to connect a 20 V zener diode between the gate and source terminals to protect against voltage transients. Such transients may come from the power supply, or may be internally transferred from the drain circuit. Once a power MOSFET switching circuit has been placed in proper operation, gate damage is unlikely to occur. Although very thin, the silicon dioxide gate insulation compares favorably with the highest quality capacitor dielectric materials. (1700)

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Snubber network values

The RC snubber can be empirically designed by starting with several hundred ohms of no inductive resistance, then determining the minimum capacitance that satisfactorily minimizes the turnoff transient; then further experimentation with R and C can optimize results. Although empirical procedure is also rewarding, it is best to first arrive at ballpark values by calculating R and C. For the capacitor, this objective is attained by means of the relationship:

С= ( I ) (t_f) / V,

where I is the peak switching current, t_f is the fall-time of the switching transistor, and V is the peak switching voltage.

The fall time, t_f , is readily available from the specifications of the device. Interestingly, algebraic rearrangement of this equation yields:

CV = ( I ) (t_f ) ,

where the quantities on both sides of the equal sign represent Q, or charge, Translated into circuit operation, the capacitor absorbs the change in charge produced when the transistor turns off. This is tantamount to stating that the turn-off transient is transferred from the switching device to the capacitor.

Note also that V tends to be approximately twice the power-supply voltage for single-transistor switching circuits. (In contrast, V is equal to the power-supply voltage when push-pull, half-bridge, and full-bridge switching circuits are used.)

Now, what about R for the snubber network? It turns out that R is a function of the minimum on time of the switching waveform, and can be determined from:

R = t_on / C .

Where t_on represents the minimum pulse-duration, or on time. Also, the power rating of the resistance, R, is given by

P = C V ( f ) / 2 .

In all of the foregoing relationships, R is expressed in ohms, C is expressed in Farads, V is expressed in volts, /is expressed in amperes, P is expressed in watts, f is expressed in Hertz, and t_f and t_on are expressed in seconds. This is the dimensional standard advocated by most textbooks on physics. (1900)

TEXT 11

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Phase-control facts

Power systems using thrusters often exert control of load power by control of the time of firing during the ac cycle. This implies delaying conduction of the thruster so that only a selectable portion of the ac cycle is available for the load. The non-sinusoidal voltage and current wave applied to the load has operational, measurement, analytic, and harmonic interference ramifications not encountered with straight dc or with ac sine-wave power. However, such phase-control circuits are relatively simple, efficient, and very convenient. Efficiency is high because turn on is extremely fast, and turn off is reasonably fast. Also, conductive losses tend to be low because of the volt or so dropped across these devices. Manufacturers provide a wide variety of thyristors, from signal-level types to giant devices capable of handling kilovolts and kilo-amperes simultaneously. A nice thing about using thyristors for phase-controlled power is the self commutating feature, wherein turn off automatically obtains when the current wave goes through zero. Phase-controlled power can be accomplished either on a half-wave or on a full-wave basis. A single SCR (silicon-controlled rectifier) yields a half-wave circuit, whereas triacs enable full-wave operation from a single device.

Several relationships pertaining to phase-count oiled power are often overlooked or become tr& sources of confusion. As might be suspected, a half-wave circuit can only deliver half the load power forthcoming from a full-wave circuit. However, it follows that the RMS (root mean square) current capability of a half wave controller is then 70.7 percent, not 50 percent of a full-wave system I same token, the maximum RMS voltage that a half-wave circuit can deliver to the load is 70.7 percent that of a full-wave controller; the full-wave technique can deliver just about the same RMS load voltage as the RMS line voltage.

There is often needless worry that sufficient control range will be attained by a phase-control system. Such apprehension is generally unfounded for the following reasons: in a full-wave control circuit, a conduction angle of 30 degrees corresponds to only 3 percent of full-load power; a conduction angle of 150 degrees provides 97 percent of full-load power. This means that 94 percent of full power control is avail­able from a phase adjustment of only 120 degrees. Effort to produce wider phase adjustment obviously leads to greatly diminishing returns. Numbers for a half-wave control situation are different, but likewise lead to the conclusion that phase-control throughout the 30- to 150-degree range suffices for practical power control. (2250)

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